Power convertor with low loss switching

ABSTRACT

A welding power supply includes an input rectifier that receives sinusoidal or alternating line voltage and provides a rectified voltage. A pre-regulator provides a dc bus and a convertor, such as a boost convertor, provides a welding output. The pre-regulator is an SVT (slow voltage transition) and an SCT (slow current transition) switched convertor. It may include a snubber circuit having a diode that is SVT switched. Also, the boost convertor may be SVT and SCT switched. The pre-regulator preferably includes a power factor correction circuit. The power source includes, in one embodiment, an inverter having a snubber circuit having a first switch in anti-parallel with a first diode, and a second switch in anti-parallel with a second diode. The first switch and first diode are connected in series with the second switch and the second diode, and the first and second switches are connected in opposing directions, to form a switched snubber.

FIELD OF THE INVENTION

[0001] This invention relates generally to power sources used in weldingand, more particularly, to welding power sources that have apre-regulator.

BACKGROUND OF THE INVENTION

[0002] Power sources typically convert a power input to a necessary ordesirable power output tailored for a specific application. In weldingapplications, power sources typically receive a high voltage,alternating current, (VAC signal and provide a high current weldingoutput signal. Around the world, utility power sources (sinusoidal linevoltages) may be 200/208 V, 230/240 V, 380/415 V, 460/480 V, 500 V and575 V. These sources may be either single-phase or three-phase andeither 50 or 60 Hz. Welding power sources receive such inputs andproduce an approximately 10-75 volt. DC or AC high current weldingoutput.

[0003] There are many types of welding power sources that provide powersuitable for welding, including inverter-based welding power sources. Asused herein, an inverter-type power supply includes at least one stagewhere DC power is inverted into ac power. There are several well knowninverter type power sources that are suitable for welding. These includeboost power sources, buck power sources, and boost-buck power sources.

[0004] Traditionally, welding power sources were designed for a specificpower input. In other words, the power source cannot provide essentiallythe same output over the various input voltages. More recently, weldingpower sources have been designed to receive any voltage over a range ofvoltages, without requiring relinking of the power supply. One prior artwelding power supply that can accept a range of input voltages isdescribed in U.S. Pat. No. 5,601,741, issued Feb. 11, 1997 to Thommes,and owned by the assignee of the present invention, and is herebyincorporated by reference.

[0005] Many prior art welding power supplies include several stages toprocess the input power into welding power. Typical stages include aninput circuit, a pre-regulator, an invertor and an output circuit thatincludes an inductor. The input circuit receives the line power,rectifies it, and transmits that power to the pre-regulator. Thepre-regulator produces a dc bus suitable for conversion. The dc bus isprovided to the invertor of one type or another, which provides thewelding output. The output inductor helps provide a stable arc.

[0006] The pre-regulator stage typically includes switches used tocontrol the power. The losses in switches can be significant in awelding power supply, particularly when they are hard switched. Thepower loss in a switch at any time is the voltage across the switchmultiplied by the current through the switch. Hard switching turn-onlosses occur when a switch turns on, with a resulting increase incurrent through the switch, and it takes a finite time for the voltageacross the switch to drop to zero. Soft switching attempts to avoidturn-on losses by providing an auxiliary or snubber circuit with aninductor in series with the switch that limits the current until thetransition to on has been completed, and the voltage across the switchis zero. This is referred to as zero-current transition (ZCT) switching.

[0007] Similarly, hard switching turn-off losses also occur when aswitch turns off, with a resultant rise in voltage across the switch,and it takes a finite time for the current through the switch to drop tozero. Soft switching attempts to avoid turn-off losses by providing anauxiliary or snubber circuit with a capacitor across the switch thatlimits the voltage across the switch until the transition to off hasbeen completed, and the current through the switch is zero. This isreferred to as zero-voltage transition (ZVT) switching.

[0008] There are numerous attempts in the prior art to providesoft-switching power converters or invertors. However, these attemptsoften either transfer the losses to other switches (or diodes) and/orrequire expensive additional components such as auxiliary switches andtheir control circuits. Thus, an effective and economical way ofrecovering (or avoiding) switching losses in power converters orinverters is desirable. Examples of various attempts at soft switchingare described below.

[0009] U.S. Pat. No. 5,477,131, issued Dec. 19, 1995 to Gegner disclosesa ZVT type commutation. However, a controlled auxiliary switch and acoupled inductor are needed to implement the ZVT. Also, the primarycurrent is discontinuous.

[0010] Some prior art designs require discontinuous conduction mode fordiode recovery. One such design is found in U.S. Pat. No. 5,414,613.This is undesirable because of excessive high frequency ripple in thepower lines.

[0011] Gegner also disclosed a ZVS converter that operated in amulti-resonant mode in U.S. Pat. No. 5,343,140. This design producedrelatively high and undesirable RMS current and RMS voltage.

[0012] Another multi-resonant converter is disclosed in U.S. Pat. No.4,857,822, issued to Tabisz. This design causes undesirable high voltagestress during ZVS events and undesirable high current stress during ZCSevents.

[0013] U.S. Pat. No. 5,307,005 also requires an auxiliary switch. Lossesoccur when the auxiliary switch is turned off. This merely shiftsswitching losses, rather than eliminating them. Other designs that“shift” losses are shown in U.S. Pat. Nos. 5,418,704 and 5,598,318.

[0014] A circuit that requires an auxiliary controlled switch but doesnot “shift” losses to the auxiliary switch is shown in U.S. Pat. No.5,313,382. This is an improvement over the prior art that shiftedlosses, but still requires an expensive controlled switch.

[0015] Another design that avoided “loss shifting” is shown in U.S. Pat.No. 5,636,144. However, that design requires a voltage clamp forrecovery spikes, and 3 separate inductors. Also, the voltages on theinductors is not well controlled.

[0016] A zero-current, resonant boost converter is disclosed in U.S.Pat. No. 5,321,348. However, this design requires relatively complexmagnetics and high RMS current in the switches and magnitudes. Also, ahigh reverse voltage is needed for the boost diodes.

[0017] When it is not practical or cost effective to use a true ZCT andZVT circuit, an approximation may be used. For example, slowvoltage/current transitions (SVT and SCT) as used herein, describetransitions where the voltage or current rise is slowed (rather thanheld to zero), while the switch turns off or on.

[0018] A typical prior art welding power supply 100 with a pre-regulator104 and an output convertor or inverter 105 is shown in FIG. 1. An inputline voltage 101 is provided to a rectifier 102 (typically comprised ofa diode bridge and at least one capacitor). Pre-regulator 104 is ahard-switched boost converter which includes a switch 106 and aninductor 107. A diode 108 allows a capacitor 109 to charge up by currentflowing in inductor 107 when the switch 106 is turned off. The currentwaveform in inductor 107 is a rectified sinusoid with high frequencymodulation (ripple).

[0019] The amount of ripple may be reduced by increasing the frequencyat which switch 106 is switched. However, as the frequency at which aprior art hard switched boost converter is switched is increased toreduce ripple, the switching losses can become intolerable.

[0020] Another drawback of some prior art power supplies is a poor powerfactor. Generally, a greater power factor allows a greater power outputfor a given current input. Also, it is generally necessary to have morepower output to weld with stick electrodes having greater diameters.Thus, a power factor correction circuit will allow a given welding powersupply to be used with greater diameter sticks for a given line power. Aprior art inverter that provided a good power factor is disclosed inU.S. Pat. Nos. 5,563,777. Many prior art convertors with power factorcorrection suffer from high switching losses. Examples of such prior artdesigns are found in U.S. Pat. Nos. 5,673,184; 5,615,101; and 5,654,880.

[0021] One type of known output convertor is a half-bridge, transformerisolated, inverter. However, such output invertors often have highswitching losses and/or require passive snubber circuits (whichincreases losses) because each snubber must operate in both directionsoverall, but only in one direction at a time. Also, known snubbercircuits generally have a limited range of acceptable loads and will notsnub proportional to the load, thus the losses are relatively high forlower loads.

[0022] Accordingly, a power circuit that provides little switchinglosses and a high (close to unity) power factor is desirable. Also, thepre-regulator should be able to receive a wide range of input voltageswithout requiring re-linking. A desirable output convertor will includea full wave, transformer isolated, inverter, that is soft switch and hasfull range, full wave, low loss snubber.

SUMMARY OF THE PRESENT INVENTION

[0023] According to a first aspect of the invention a welding powersupply includes an input rectifier that receives sinusoidal oralternating line voltage and provides a rectified sinusoidal voltage. Apre-regulator receives the rectified input-and provides a dc bus. Aninvertor connected across the bus provides a welding output. Thepre-regulator is an SVT (slow voltage transition) and an SCT (slowcurrent transition) switched invertor.

[0024] In one embodiment the pre-regulator includes a snubber circuithaving a diode that is SVT switched.

[0025] In another embodiment the inverter is a boost converter with aswitch. The pre-regulator includes a snubber circuit having a capacitorand an inductor. The capacitor is connected to slow the switch voltagerise while the switch is turning off, and the inductor is connected toslow the switch current rise when the switch is turning on. The boostconverter includes a boost inductor, a switch, and an output capacitorin another embodiment. Also, the snubber includes a snubber capacitor, asnubber inductor, a first snubber diode, a second snubber diode, a thirdsnubber diode, a fourth snubber diode, and first and second snubbercapacitors. The snubber inductor, switch, and fourth diode are connectedsuch that current may flow from the boost inductor to any of the snubberinductor, switch, and fourth diode. Current flowing through the fourthdiode can flow through either the third diode or the second capacitor.Current flowing from the boost inductor through the snubber inductor canflow through either the first diode or the first capacitor. The fourthdiode and the second capacitor are connected across the switch andcurrent flowing through the third diode can flow through either thefirst capacitor and the snubber inductor or through the second diode.Current flowing through the fist and second diodes flows to the output.A fifth diode is connected in anti-parallel to the switch in oneembodiment.

[0026] A second aspect of the invention is a method of providing weldingpower by rectifying a sinusoidal or alternating input line voltage andpre-regulating the sinusoidal input line voltage to provide a dc bus.The method further includes SVT and SCT switching a boost convertor. Thebus is converted to a welding output.

[0027] Pre-regulating includes, in one embodiment, maintaining a boostconverter switch off, and allowing current to flow through a boostinductor, a snubber inductor, and a first diode, to the dc bus, andturning the switch on and diverting current from the snubber inductor tothe switch. Current is reversed in the snubber inductor and a secondcapacitor is discharged through a first capacitor, a third diode, andthe snubber inductor, thereby transferring energy from the secondcapacitor to the snubber inductor. Current is diverted through a fourthdiode, the third diode and the first capacitor when the second capacitoris discharged, thereby transferring energy from the snubber inductor tothe first capacitor. The switch is turned off and current divertedthrough the fourth diode and into the second capacitor. Voltage on thesecond capacitor is allowed to rise until current begins to flow fromthe snubber inductor to the first capacitor and then current is divertedfrom the second capacitor through a third diode to the second diode. Thecurrent from the boost inductor to the snubber inductor increases untilall of the current from the boost inductor flows into the snubberinductor. Then current is diverted from the first capacitor to the firstdiode. This process is repeated.

[0028] One embodiment includes SVT turning off a diode in a snubbercircuit. Another includes slowing the switch voltage with a capacitorrise while the switch is turning off, and slowing the switch currentrise with an inductor while the switch is turning on to SVT and SCTswitching a boost convertor.

[0029] A third aspect of the invention is a welding power supply havingan input rectifier that provides a rectified voltage. A pre-regulatorreceives as an input the rectified signal and provides a dc bus. Aninvertor converts the bus to a welding output and the pre-regulatorincludes a power factor correction circuit.

[0030] Yet another aspect of the invention is a welding power supplyhaving an input rectifier and a preregulator and an invertor. Thepre-regulator includes a snubber circuit having a first switch inanti-parallel with a first diode, and a second switch in anti-parallelwith a second diode. The combination of the first switch and first diodeare connected in series with the combination of the second switch andthe second diode, and the first and second switches are connected inopposing directions.

[0031] Another aspect of the invention is a welding power supply havingan inverter with first and second current paths through a transformer,each in a unique direction. The first current path includes at least afirst switch with an anti-parallel first diode and the second currentpath through the transformer in a second direction, the second currentpath including at least a second switch with an anti-parallel seconddiode. A snubber includes a current path having a third switch with ananti-parallel third diode, a fourth switch with an anti-parallel fourthdiode. The third switch and anti-parallel diode are in series with, andoppositely directed from, the fourth switch and anti-parallel diode. Thesnubber also has a at least one snubber capacitor.

[0032] In alternative embodiments the first and second switches are in ahalf-bridge configuration or full bridge configuration. Also, thesnubber capacitor may be split into two capacitors.

[0033] Another aspect of the invention is a method of providing weldingpower by turning on a first power switch and a first snubber switch, andallowing current to flow through the first power switch, a first dc bus,a first power capacitor, and in a first direction through a transformer.Then the first power switch is turned off and current flows through thefirst snubber switch, a second snubber diode, a snubber capacitor, andthrough the transformer in the first direction, while the first powerswitch is turning off, to provide a slow voltage transition off. Thencurrent flows through a second anti-parallel power diode, a second DCbus, a second power capacitor, and through the transformer in the firstdirection, while the first power switch is continuing to turn off, tocontinue providing a slow voltage transition off. The first snubberswitch is also turned off. After the system is at rest a second powerswitch on and a second snubber switch are turned after the first powerswitch is off, and current flows through the second power switch, thetransformer in a second direction, the second power capacitor, and thesecond bus. The second power switch is turned off and current flowsthrough the second snubber switch, a first snubber diode, thetransformer in the second direction, and a snubber capacitor, while thesecond power switch is turning off, to provide a slow voltage transitionoff. Then current flows through a first power diode, the transformer inthe second direction, and the first power capacitor, while the secondpower switch is turning off, to provide a slow voltage transition off.The second snubber switch is also turned off, and the process isrepeated.

[0034] Other principal features and advantages of the invention willbecome apparent to those skilled in the art upon review of the followingdrawings, the detailed description and the appended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

[0035]FIG. 1 is a circuit diagram of a prior art welding power supplyhaving a boost convertor pre-regulator;

[0036]FIG. 2 is a block diagram of a welding power supply constructed inaccordance with the present invention;

[0037]FIG. 3 is a circuit diagram of a power factor correction circuitused in the preferred embodiment;

[0038]FIG. 4 is a circuit diagram of the pre-regulator of FIG. 2;

[0039] FIGS. 5-13 are the circuit diagram of FIG. 4 showing variouscurrent paths;

[0040]FIG. 14 is a circuit diagram of a switching circuit;

[0041]FIG. 15 is a full wave inverter using the switching circuit ofFIG. 14;

[0042]FIG. 16 is a control circuit; diagram; and

[0043] FIGS. 17-22 are the circuit diagrams of FIG. 15 showing variouscurrent path.

[0044] Before explaining at least one embodiment of the invention indetail it is to be understood that the invention is not limited in itsapplication to the details of construction and the arrangement of thecomponents set forth in the following description or illustrated in thedrawings. The invention is capable of other embodiments or of beingpracticed or carried out in various ways. Also, it is to be understoodthat the phraseology and terminology employed herein is for the purposeof description and should not be regarded as limiting. Like referencenumerals are used to indicate like components.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0045] While the present invention will be illustrated with reference toa welding power supply using a boost converter for a pre-regulator andparticular circuitry, it should be understood at the outset that othercircuit topologies may be used, and the power supply may be used forother purposes, and still be within the intended scope of thisinvention.

[0046] A block diagram of a welding power supply constructed inaccordance with the preferred embodiment is shown in FIG. 2. Source 201represents the input line voltage used to provide power to the weldingpower supply. The input line voltage may be anywhere between 90 and 250volts in the preferred embodiment. The voltage typically operates at afrequency of 60 hertz (in the United States) and is single phase in thepreferred embodiment (although alternative embodiments use a three phaseinput). Other voltages may also be used.

[0047] The input voltage is provided to a rectifier 202, which may be asimple bridge rectifier. The output of rectifier 202 is a rectifiedsinusoid.

[0048] A pre-regulator 204 receives the rectified sinusoid fromrectifier 102 and provides a dc bus output to an output invertor 205.Pre-regulator 204, in the preferred embodiment is a soft-switched boostconvertor which provides close to a unity power factor. Other convertorpr invertor configurations may be used. Pre-regulator 204 also allowsthe input voltage to be anywhere within a range of input voltages in thepreferred embodiment.

[0049] Convertor 205 is preferably a half-bridge, transformer isolated,soft (or slow) switched invertor. Such an output circuit will bedescribed in detail below. Output convertor 205 may alternatively be atypical forward convertor (generally a buck convertor and atransformer), and other output converters may be used in otherembodiments. A circuit including an output buck convertor is describedin a U.S. Patent Application entitled Auxiliary Open Circuit VoltagePower Supply, invented by Vogel and Geissler, filed on even dateherewith, (hereby incorporated by reference) and assigned to theassignee of this invention. The output of convertor 205 is providedthrough an inductor 207 to welding output 208.

[0050] The circuit used in the-preferred embodiment to implementpre-regulator 204 is shown in FIG. 4 (along with rectifier 202 andvoltage source 201). The embodiment of FIG. 4 uses a 90-250 volt acpower line as input voltage 201. Rectifier 202 is comprised of diodesD6, D7, D8, and D9, which rectify the input voltage to provide a singlepolarity sinusoidal input voltage.

[0051] The power factor correction portion (described below) ofpre-regulator 204 functions best when the input voltage is sinusoidal,although it could be another alternating input. Thus, a small (10 μF)capacitor (not shown) is provided across input rectifier 202 in oneembodiment to smooth the input line voltage.

[0052] The rectified input voltage is applied to a boost inductor L1(750 μH) which is connected with a boost switch Z1 (preferably an IGBT)to form a boost convertor. An anti-parallel diode D5 is connected acrossswitch Z1 to protect switch Z1 during transitions. The portion of thecircuit which provides the lossless switching includes a snubberinductor L2 (3.9 μH) a pair of capacitors C1 (1 μF) and C2 (0.068 μF),and diodes D1, D2, D3, and D4. Switch Z1 is switched in a known mannersuch that the output of pre-regulator 204 is a desired voltage, nomatter what the input voltage is, The output is provided across acapacitor C5 (2000 μF) that provides a stable voltage source (400 voltsin the preferred embodiment) for the downstream convertor. Also,capacitor C5 prevents the voltage from being dangerously high anddamaging switch Z1.

[0053] The soft switching of pre-regulator 204 is best understood withreference to FIG. 5-11, which show the circuit with various currentpaths (states). The first state (FIG. 5) is when switch Z1 is off, andthe current (arrow 501) is in a steady state condition through inductorsL1 and L2, and diode D1, to charge output capacitor C5 (arrow 501).

[0054] Then, switch Z1 is turned on, and current from inductor L1 beginsto be directed through switch Z1 (arrow 601 of FIG. 6). Switch Z1applies a reverse voltage to inductor L2, causing its current to fall.Thus, the current (in this state) is decreasing through inductor L2 andincreasing through switch Z1. Inductor L2 effectively limits or slowsthe current in switch Z1 at turn on until the switch voltage drops (toclose to zero). Thus, the turn on has been a slow-current transition(SCT).

[0055] Eventually all of the current from inductor L1 flows throughswitch Z1, and current in inductor L2 drops until it becomes zero, andthen reverses. Capacitor C2 discharges through capacitor C1, diode D3,and inductor L2, as shown in FIG. 7, by arrow 701. Capacitors C1 and C2allow diode D1 to turn off with a SVT, thus reducing losses. Thedischarge occurs at a resonant frequency determined by the time constantof the inductance of inductor L2 and the series capacitance ofcapacitors C1 and C2 (f=1/(2n{square root} (L2*(C1+C2/C1*C2)). The timeit takes for capacitor C2 to discharge is the SVT time for diode D1.

[0056] Capacitor C2 discharges to about zero volts, and diode D4 beginsto conduct, as shown by arrow 801 in FIG. 8. When diode D4 conducts,inductor L2 releases the energy stored therein to capacitor C1 at aresonant frequency determined by inductor L2 and capacitor C1(f=1/(2n{square root} (L2*C1))). The voltage energy on capacitor C1 istransferred to current in inductor L2, and then to voltage on capacitorC1. The ratio of voltage transfer is nearly equal to the capacitanceratio.

[0057] When the charge transfer is complete, and current ceases to flowin snubber inductor L2, the snubber is reset, and current in inductor L1increases through switch Z1, as shown in FIG. 9. The circuit remains inthis state until the switch is turned off.

[0058] Next, switch Z1 is turned off, (FIG. 10) and current is divertedthrough diode D4 and into capacitor C2 (arrow 1001). Capacitor C2provides the SVT time for switch Z1, thus a soft switching off isprovided. The voltage on capacitor C2 continues to rise and eventuallyreaches the bus voltage (the voltage on capacitor C5) less the voltageon capacitor C1.

[0059] When this happens the voltage on capacitor C1 begins toreestablish the current in inductor L2 (FIG. 11 and arrow 1101). Thevoltage on capacitor C2 continues to rise until it reaches the busvoltage plus two diode drops. At that time current from inductor L1 nottaken by inductor L2 is diverted through diode D3 FIG. 12 and arrow1201). The voltage on capacitor C1 continues to increase the current ininductor L2.

[0060] Eventually all of the current from inductor L1 flows throughinductor L2, and current through diodes D3 and D4 ceases (FIG. 13).Capacitor C1 continues to give energy to the bus.

[0061] When all of the energy-on capacitor C1 is expended (to the bus)current flows from inductor L1 to inductor L2, and through diode D1.This is the state initially described, with respect to FIG. 5, and thecycle repeats.

[0062] Thus, the voltage rise across switch Z1 was slowed by capacitorC2 to allow the current to drop when switch Z1 was turned off. Thecurrent rise in switch Z1 was slowed by inductor L2 to allow the voltageto drop, when switch Z1 was turned on. Moreover, diode D1 wassoft-switched by capacitors C1 and C2.

[0063] The portion of pre-regulator 204 that provides power factorcorrection is a power factor correction circuit 404 (FIG. 4), andgenerally senses the input voltage waveform, and conforms the shape ofthe current waveform to be that of the line voltage waveform. Thisprovides a power factor of very close to 1, 0.99 in the preferredembodiment. Power factor correction circuit 404 may be implemented usingan integrated circuit, such as a UC3854 or an ML4831, or with discretecomponents. Power factor correction circuit 404 receives as inputs theoutput voltage from rectifier 202, the output voltage from pre-regulator204, and the output current of pre-regulator 204 (using a CT 405).Because the frequency of pre-regulator 204 (25 KHz) is much higher thanthat of the line (60 Hz) the pre-regulator current can be made to trackthe input line voltage shape by sensing the shape of the input voltage,and controlling the input current in response thereto.

[0064] An embodiment of power factor correction circuit 404 havingdiscreet components is shown in FIG. 3 controls the switches so that theinput current is shaped to match the input voltage, as well as regulatesthe DC bus.

[0065] The input voltage is rectified and provided to a Z pole Besselfilter which removes switching frequencies. We Bessel filter includescapacitors 1602 (0.0022 μF) and 1603 (0.001 μF), resistors (1606-1608(1M ohms), resistors 1609-1610 (39.2 K ohm), and an op amp 1615. Theoutput of the Bessel filter (V-RECT) is provided to a low pass filter(approximately 2 Hz), which includes resistors 1611 and 1612 (68.1 Kohms), capacitor 1604 (0.22 μF), capacitor 1605 147 μF), and op amp1616. The output of op amp 1616 gives an average of the input linevoltage (V-LINE).

[0066] V-LINE is provided to a typical precharge circuit 1625 which setsa delay before the electrolytic capacitors in the power supplyprecharge. An op amp 1626, and resistors 1629 (100 K ohms) and 1630 (10K ohms) don't allow a capacitor 1627 (10 μF) to charge through aresistor 1628 (100 ohms) until the line voltage reaches a threshold.After the line voltage reaches the threshold capacitor 1627 charges to alevel where it turns on a relay (not shown) through associatedcomponents including a resistor R63 (200 K ohms), a resistor R51 (100 Kohms), a resistor R108 (619 K ohms), an op amp U1, a diode D57, a NANDgate U2, and a resistor R89 (4.7 K ohm). These components operate in atypical fashion. The relay energizes and fires an SCR that prechargesthe electrolytic capacitors.

[0067] A multipliers/divider 1631 of receives the rectified sine voltagesignal, and divides that by the average input (typically either 230 or460) so that a scaled rectified voltage is provided. Then the scaled, isrectified voltage is multiplied by an error signal from the bus toproduce a reference command. Specifically, V-RECT, the output of op amp1615 which corresponds to the rectified input voltage, is providedthrough a resistor 1632 (100 K ohms) and an op amp 1633 as one channelinput to the multiplication. The other channel input to themultiplication is a BUS-ERROR signal provided through an op amp 1636A.

[0068] The output of op amp 1633 is provided through a log transistor1635, and the average line voltage (V-LINE) is provided through an opamp 1636 and to a log transistor 1637. The common junction betweentransistors 1635 and 1637 is a subtraction, so the base of transistor1637 is the result of the subtraction. That difference is added througha transistor 1638 to the bus error. The sum is provided to a transistor1639, which takes the anti-log of that value. Thus, a division andmultiplication are performed. The output is scaled by an op amp 1641 andassociated circuitry including a diode 1642, a capacitor 1643 (0.001μF), and a resistor 1645 (20 K ohms).

[0069] A transistor 1646 limits the output current of op amp 1641, andis controlled by a resistor 1647 (20K ohms) and a diode 1642. The inputto op amp 1650 is a scaled bus voltage, and sets the maximum outputcommand. The output command (VCOMM) is used to force the current shapeto match the input voltage shape.

[0070] The BUS-ERROR signal is provided by a typical error circuit whichincludes an op amp 1651 and associated circuitry resistors 1653 (20Kohms), 1654 (11K ohms), and 1655 (499K ohms), a diode 1657 and acapacitor 1658 (0.047 μF). An 8 volt reference signal is compared to thedivided down (and scaled) 800 volt bus. An error signal is providedthrough a resistor 1659 (82.5K ohms) to op amp 1636A commanding anincrease or decrease in the current to increase or decrease the busvoltage. Also, the current command is adjusted by the shape of the inputsignal as provided through V-RECT to mimic the shape of input rectifiedsignal. Therefore, the current needed to result in a desired bus voltageis provided, but in such a shape that a power factor very close to oneis obtained.

[0071] The command signal is summed with a current feedback signal froma CT1 by an op amp 1670 and provided to be a boost drive circuit 1660through logic gates (not shown) to turn on and off the IGBT in thepreregulator. A CT is used to provide current feedback (rather than anLEM for example) because if a LEM fails it will call for unlimitedcurrent.

[0072] The boost drive signal is a digital signal of either zero (IGBTON) or fifteen volts (IGBT OFF). The boost drive input is provided tothe base of a pair of transistors 1661 and 1662 because the logic gatesoutput do not provide enough current to drive the IGBT's. Thus,transistors 1661 and 1662 provide sufficient current. A transistor 1663level shifts. The gates of a pair of transistors 1665 and 1666 are tiedtogether by a capacitor 1667 (0.1 μF).

[0073] Another aspect of this invention is implemented with ahalf-bridge, transformer isolated, inverter that is SVT switched. Theinvertor uses a switch circuit 1400, shown in FIG. 14, that includes apair of switches or IGBT's 1402 and 1403, and a pair of diodes 1404 and1405. Diode 1404 is an anti-parallel diode for switch 1402. Diode 1405is an anti-parallel diode for switch 1403. The two switch/diode parallelcombinations are in series, but reversed, i.e. in opposing directions.This configuration provides a diode-type switch whose direction can bereversed.

[0074] An invertor using switch circuit 1400 is shown in FIG. 15, andincudes a dc voltage source 1501, a pair of switches 1502 and 1504, witha pair of anti-parallel diodes 1503 and 1505, a pair of capacitors 1507and 1508 (1410 μF), a transformer 1509, a capacitor 1512 (0.099 μF), anoutput rectifier including diodes 1510 and 1511, and an output inductor1513.

[0075] Capacitor 1512 is switched across transformer 1509 by switches1502 and 1504. Switches 1402 and 1403 are used to soft switch switches1502 and 1504. Switches 1402 and 1403 do not need any special timing,and run with the main clock at effectively 50% duty cycle. For example,switches 1502 and 1402 turn on together, and switch 1502 deliverscurrent to transformer 1509, while switch 1402 does nothing. When switch1502 turns off, switch 1402 remains on, and current is directed throughswitch 1402 and diode 1405 into capacitor 1512, thus giving an SVT (SlowVoltage Transition) turn off. Switch 1402 is turned off after thetransition and diode 1405 prevents the back flow of current fromcapacitor 1512. This occurs in complimentary fashion with switches 1501and 1402 and diode 1405. Thus, this circuit provides full-wavetransformer usage, PWM control, complete capacitor balance control withno extra circuitry, and efficient use of switches with SVT.

[0076] Referring now to FIGS. 17-24, the various current paths followedduring a complete cycle are shown. The circuit in these Figures is analternative embodiment that includes splitting capacitor 1512 into twocapacitors, one connected to the upper bus, and one connected to thelower bus. This is done because the path through capacitors 1507 and1508 can reduce the effectiveness of the snubber substantially.

[0077] Initially, all of the switches are off in the snubber, andcapacitors 1512A and 1512B split the bus. The 800 volts bus is alsosplit by capacitors 1507 and 1508 (for half bridge operation).Capacitors 1507 and 1508 should be large enough to keep the junctionvoltage between them substantially constant during operation. Switches1504 and 1402 are turned on together. Switch 1504 delivers power totransformer 1509 while switch 1402 is blocked by diode 1404. Thus,switch 1402 is in “standby” until switch 1504 turned off, as shown inFIG. 17.

[0078] Switch 1504 turns off, and the current through the transformertransfers to switch 1403, diode 1404 and capacitor 1512 (which form thesnubber path). The voltage across switch 1504 rises slowly giving a slowvoltage transition. This current path is shown in FIG. 18.

[0079] When the voltage across switch 1504 reaches the bus voltage, theremaining energy from transformer 1509 is spilled back into the busthrough diode 1503. This current path is shown in FIG. 19. When theremaining energy and transformer 1509 has been provided to the bus thesystem comes to rest with snubber capacitor 1512 fully charged tocompletely soft switch 1502 (FIG. 15).

[0080] After the system comes to rest switches 1502 and 1403 are turnedon together. Switch 1502 delivers power to transformer 1509 while switch1403 is blocked by diode 1404. Thus, switch 1403 is in standby untilswitch 1502 is turned off (FIG. 20).

[0081] Switch 1502 is turned off and current from transformer 1509transfers to capacitor 1512 through the snubber path including diode1404 and switch 1403 rises slowly, giving a slow voltage transition.This current path is shown in FIG. 21. The current continues in thispath until the voltage across switch 1502 has reached the bus, andremaining energy in transformer 1509 is spilled into the bus throughdiode 1505 (FIG. 22). Capacitor 1512 fully charged so that switch 1504may be soft switched. The process then repeats.

[0082] One feature of the switched snubber used in FIGS. 15-22 is thatthe main switches (1504 and 1502) do not incur actual losses if theoutput power is less than necessary to transition snubber capacitor 1512from “rail to rail”. Thus, it is not necessary to fully transition thesnubber. The reversible single direction switch prevents snubberinterference on turn on, and thus provides snubbing proportional toload. This feature allows very heavy snubbing without restricting theload range of the inverter.

[0083] An alternative embodiment includes using a full bridge version ofthe snubber.

[0084]FIG. 16 shows a control circuit for controlling the switching ofthe switched snubber in FIGS. 14-22. Four gate drives 1402A-1405A are,used to provide the gate signals for switches 1402-1405, respectively.These gate drives are not shown in detail and are conventional gatedrives such as those found in the Miller XMT 304®. The gate drives areinverting in that a high output maintains the gates off and a low outputmaintains the gates off.

[0085] Gate drivers 1402A-1405A are controlled by a logic circuit 2301.Logic circuit 2301 includes a plurality of NAND and OR gates in thepreferred embodiment, however it's specific construction may be any ofthe designer's choosing. And enable signal is included as an input tologic circuit 2301, in one embodiment. They enable signal is used onlyduring power down.

[0086] An error amplifying circuit 2303 is also shown in FIG. 16. Erroramplifying circuit 2303 may be a standard error circuit and is used, inthe preferred embodiment, with a CT feedback signal. The output of erroramplifier circuit 2303 is a PWM reference signal. The PWM referencesignal control is provided through an opto-isolator 2305 to electricallyisolate the remaining portion of the circuit from the error amp circuit.A pair of resistors 2306 (10K ohms) and 2307 (2K ohms) scale the PWMreference command for input into opto-isolator 2305. The output ofopto-isolator 2305 is scaled from a current to a voltage by a resistor2308 (10K ohms).

[0087] Generally, the control circuitry implements a modified PWMcontrol scheme. Above a minimum pulse width operation is a typical PWMscheme, and the pulse width is adjusted to increase or decrease current.However, for current less than that corresponding to the minimum pulsewidth the frequency of pulses is reduced (thus increasing the OFF time).The minimum pulse width is used because the gate drives have a limitedspeed.

[0088] The conventional pulse with modulation portion works with a rampcreated by an op amp 2310, resistors 2311 (10K ohms), 2312 (10K ohms),and 2313 (200K ohms). The PWM reference command is received by op amp2310 through a diode 2314. The appropriate switch is turned on at thestart of the ramp. The ramp is initiated by an op amp 2315 and resistors2316 (10K ohms), 2317 (611 ohms), 2318 (20K ohms), 2319 (200K ohms),2321 (6.11K ohms) and 2322 (2K ohms).

[0089] The main power switches (1502 and 1504) are maintained on for 95%of the total ramp time. The 95% threshold is set by an op amp 2325 and aresistor 2326 (10K ohms). The switches are turned off by changing stateson the set input of a flip flop 2327 (which is connected to op amp2325). The snubber switches (1402 and 1403) are switched off at 100% ofthe ramp.

[0090] A current source including transistors 2330 and 2331 andresistors 2333 (332 ohms), 2334 (100 ohms) and 2335 9100 ohms). Thecurrent source sets the slop of the ramp. When a capacitor 2337 (100 pF)discharges to a threshold set by a diode 2338 the ramp is restarted. Theramp will continue up at the slope set by the current source until thecapacitor voltage reaches the threshold set by op amp 2315 and itscircuitry.

[0091] A flip flop 2328 is used to alternate between switches, and toreceive an enable signal and a machine on/off signal.

[0092] Generally, the circuit operates as follows: the capacitor voltageis fully reset down to the minimum and then the ramp begins to ramp upand the voltage on the capacitor is increasing from the current source.As the capacitor charges the output of the opto-isolator is provided toop amp 2310, which pulse width modulates the switches. When thecapacitor voltage rises above the reference voltage set by theopto-isolator, op amp 2310 changes state, causing the switch to beturned off. Steering flip flop 2328 determines which one, and only one,of the main power switches are on in a conventional manner. If thecapacitor voltage increases to the level set by op amp 2315 (95% of thepeak), then the main power switch that is on is turned off.

[0093] The frequency adjust (for low current commands) operates asfollows: the output of op amp 2315 (the ramp reset) is fed back througha NAND gate 2341 through a resistor 2342 (100K ohms) and a bufferingtransistor 2343. The machine on/off signal is also provided totransistor 2343. The output of NAND gate 2341 also causes flip flop 2328to change state through the clock input.

[0094] A voltage divider including resistors 2342 and 2345 (68.1K ohms)is tied to a diode 2346. If diode 2346 pulls down the voltage at one endof resistor 2345, then the voltage across a resistor 2347 (5.11K ohms)is also pulled down. A current mirror including resistors 2349 and 2350(100 ohms) and transistors 2351 and 2352 provide the rest current forthe ramp. However, if the voltage through 2346 is low enough, then thevoltage input to transistor 2343 will be ground, and transistor 2343will not provide current to the current mirror to reset capacitor 2333,thus allowing the ramp to continue upward.

[0095] The various aspects of this invention, while described in thecontext of a welding power supply has applications in many differentareas. Generally, in applications where low loss switching is desirableusing a boost convertor this arrangement may be used.

[0096] Numerous modifications may be made to the present invention whichstill fall within the intended scope hereof. Thus, it should be apparentthat there has been provided in accordance with the present invention amethod and apparatus for providing power with a high power factor andlow switching losses that fully satisfies the objectives and advantagesset forth above. Although the invention has been described inconjunction with specific embodiments thereof, it is evident that manyalternatives, modifications and variations will be apparent to thoseskilled in the art. Accordingly, it is intended to embrace all suchalternatives, modifications and variations that fall within the spiritand broad scope of the appended claims.

The embodiments of the invention in which an exclusive property orprivilege is claimed are defined as follows:
 1. A welding power supplycomprising: an input rectifier configured to receive an input linevoltage and provide a rectified voltage on an output; a pre-regulatorconnected to receive as an input the output of the rectifier and providea dc bus as an output; and a convertor, connected to receive the outputof the pre-regulator and provide a welding output; wherein thepre-regulator is an SVT and a SCT switched convertor.
 2. The powersupply of claim 1 , wherein the pre-regulator includes a snubber circuithaving a diode that is SCT switched.
 3. The power supply of claim 2 ,wherein the pre-regulator diode is SVT switched.
 4. The power supply ofclaim 1 , wherein the converter is a boost convertor including a switch,and the pre-regulator includes a snubber circuit having a capacitor andan inductor, wherein the capacitor is connected to slow the switchvoltage rise while the switch is turning off, and the inductor isconnected to slow the switch current rise when the switch is turning on.5. The power supply of claim 1 , wherein: the boost converter includes aboost inductor, a switch, and an output capacitor; the converterincludes a snubber, including a snubber capacitor, a snubber inductor, afirst snubber diode, a second snubber diode, a third snubber diode, afourth snubber diode, and first and second snubber capacitors; thesnubber inductor, switch, and fourth diode are connected such thatcurrent may flow from the boost inductor to any of the snubber inductor,switch, and fourth diode; current flowing through the fourth diode canflow through either the third diode or the second capacitor; currentflowing from the boost inductor through the snubber inductor can flowthrough either the first diode or the first capacitor; the fourth diodeand the second capacitor are connected across the switch; currentflowing through the third diode can flow through either the firstcapacitor and the snubber inductor or through the second diode; andcurrent flowing through the first and second diodes flows to the output.6. The power supply of claim 4 further including a fifth snubber diodeconnected in anti-parallel to the switch.
 7. A method of providingwelding power, comprising the steps of: rectifying an input linevoltage; pre-regulating the input line voltage to provide a dc bus; andconverting the dc bus to a welding output; wherein the step ofpre-regulating includes SVT and SCT switching a boost convertor.
 8. Themethod of claim 7 , wherein the step of pre-regulating includes thesteps of: maintaining a boost converter switch off, and allowing currentto flow through a boost inductor, a snubber inductor, and a first diode,to the dc bus; turning the switch on and diverting current from thesnubber inductor to the switch; reversing the current in the snubberinductor; discharging a second capacitor through a first capacitor, athird diode, and the snubber inductor, thereby transferring energy fromthe second capacitor to the snubber inductor; diverting current througha fourth diode, the third diode and the first capacitor when the secondcapacitor is discharged, thereby transferring energy from the snubberinductor to the first capacitor; turning the switch off and divertingcurrent through the fourth diode and into the second capacitor; allowingthe voltage on the second capacitor to rise until current begins to flowfrom the snubber inductor to the first capacitor; diverting current fromthe second capacitor through a third diode to the second diode; allowingthe current flowing from the boost inductor to the snubber inductor toincrease until all of the current from the boost inductor flows into thesnubber inductor; diverting current from the first capacitor to thefirst diode; and repeating these steps.
 9. The method of claim 7 ,further including the step of SVT turning off a diode in a snubbercircuit.
 10. The method of claim 9 , wherein the step of SVT and SCTswitching a boost convertor includes slowing the switch voltage risewith a capacitor while the switch is turning off, and slowing the switchcurrent rise with an inductor while the switch is turning on.
 11. Awelding power supply comprising: an input rectifier means for receivingan input line voltage and providing a rectified voltage; a pre-regulatormeans for receiving the rectified voltage and providing a dc bus,wherein the pre-regulator means is connected to the rectifier means; anda convertor means for receiving the output of the pre-regulator meansand provide a welding output, wherein the converter means is connectedto the pre-regulator means; wherein the pre-regulator means includes SVTand SCT switching means.
 12. The power supply of claim 11 , wherein thepre-regulator means includes a snubber means having a diode that is SVTswitched.
 13. The power supply of claim 11 , wherein the boost converterincludes a switch, and the pre-regulator includes a snubber circuitmeans for providing the SVT and SCT switching.
 14. A welding powersupply comprising: an input rectifier configured to receive an inputline voltage and provide a rectified voltage on an output; apre-regulator connected to receive as an input the output of therectifier and provide a dc bus as an output; and an invertor, connectedto receive the output of the pre-regulator and provide a welding output;wherein the inverter includes a snubber circuit having a first switch inanti-parallel with a first diode, and a second switch in anti-parallelwith a second diode, and wherein the combination of the first switch andfirst diode are connected in series with the combination of the secondswitch and the second diode, and wherein the first and second switchesare connected in opposing directions.
 15. A welding power supplycomprising: a first current path through a transformer in a firstdirection, the first current path including at least a first switch withan anti-parallel first diode; a second current path through thetransformer in a second direction, the second current path including atleast a second switch with an anti-parallel second diode; a snubber,including a current path having a third switch with an anti-parallelthird diode, a fourth switch with an anti-parallel fourth diode, whereinthe third switch and anti-parallel diode are in series with, andoppositely directed from, the fourth switch and anti-parallel diode, andat least one snubber capacitor.
 16. The power supply of claim 15 whereinthe first and second switches are in a half-bridge configuration. 17.The power supply of claim 15 wherein the at least one snubber capacitorincludes a first snubber capacitor connected with a first bus line and asecond snubber capacitor connected with a second bus line.
 18. The powersupply of 15 wherein the at least one snubber capacitor is in serieswith the third and fourth switches and anti-parallel diodes.
 19. Amethod of providing welding power comprising the steps of: turning on afirst power switch and a first snubber switch, and allowing current toflow through the first power switch, a first dc bus, a first powerscapacitor, and in a first direction through a transformer; turning thefirst power switch off and allowing current to flow through the firstsnubber switch, a second snubber diode, a snubber capacitor, and throughthe transformer in the first direction, while the first power switch isturning off, to provide a slow voltage transition off; allowing currentto flow through a second anti-parallel power diode, a second DC bus, asecond power capacitor, and through the transformer in the firstdirection, while the first power switch is continuing to turn off, tocontinue providing a slow voltage transition off; turning off the firstsnubber switch; turning on a second power switch and a second snubberswitch after the first power switch is off, and allowing current to flowthrough the second power switch, the transformer in a second direction,the second power capacitor, and the second bus; turning the second powerswitch off and allowing current to flow through the second snubberswitch, a first snubber diode, the transformer in the second direction,and a snubber capacitor, while the second power switch is turning off,to provide a slow voltage transition off; allowing current to flowthrough a first power diode, the transformer in the second direction,and the first power capacitor, while the second power switch is turningoff, to provide a slow voltage transition off; turning off the secondsnubber switch; and repeating these steps.
 20. The method of claim 19 ,wherein the steps of turning on the power switches includes softswitching on the power switches.
 21. The method of claim 19 wherein thestep of allowing current to flow through the second snubber diode andthe snubber capacitor includes allowing current to flow through thesecond snubber diode and a first snubber capacitor and further includesallowing current to flow through the second bus, and the secondcapacitor.
 22. The method of claim 21 wherein the step of allowingcurrent to flow through the snubber capacitor and the second snubberdiode includes allowing current to flow through the second snubber diodeand a second snubber capacitor and further includes allowing current toflow through the first bus, and the first capacitor.